Method and system for switching between different types of operation of a sensorless permanent magnet brushless motor at low or zero speed to determine rotor position

ABSTRACT

The system and method disclose for the controlling of motor switching. The system includes a controller unit having a control signal generator, a memory device, a processing unit, a signal acquisition device, and an analog-to-digital converter. A power stage has a plurality of switches and receives a control signal from the control signal generator and a power signal from a power source. The power stage drives two windings of the set of three stator windings with a multi-state pulse and leaves one stator of the three stator windings undriven. The processing unit acquires a demodulated measured voltage on the undriven winding. The processing unit communicates with the power stage to change which two windings of the three stator windings are driven when the demodulated measured voltage surpasses a threshold.

CROSS-REFERENCE

This application is a Continuation in Part to U.S. Application entitled“Circuit and Method for Sensorless Control of a Permanent MagnetBrushless Motor During Start-up,” having Ser. No. 13/800,327 filed Mar.13, 2013 and claims the benefit of U.S. Provisional Applicationentitled, “Circuit and Method for Sensorless Control of a BrushlessMotor During Start-up,” having Ser. No. 61/651,736, filed May 25, 2012,which is entirely incorporated herein by reference.

FIELD

The present disclosure is generally related to motor controllers, andmore particularly is related to a system and method for sensorlesscontrol of a permanent magnet brushless motor during start-up.

BACKGROUND

Sensored brushless motor technology is well-known and is useful forminimal flaw control at low speeds and reliable rotation. A sensoredsystem has one or more sensors that continuously communicate with amotor controller, indicated to it what position the rotor is in, howfast it is turning, and whether it is going forward or reverse. Sensorsin a sensored system increase cost and provide additional pieces thatcan break or wear down, adding durability and reliability issues.Sensorless systems can read pulses of current in the power connectionsto determine rotation and speed. Sensorless systems tend to be capableof controlling motors at higher speeds (e.g., revolutions per minute(“RPM”)), but may suffer jitter under a load at very low startingspeeds, resulting in a performance inferior to sensored brushlessmotors.

Jitter is a phenomenon that occurs with sensorless brushless motorsystems at initial starting speed and generally no longer exists afterthe motor has gained sufficient speed. Jitter comes about because at lowor zero speed, the sensorless algorithm does not have enough informationto decide which windings to energize and in what sequence. One commonsolution for starting a sensorless system is to energize one windingpair to lock the rotor at a known position. The motor windings are thencommutated at a pre-defined rate and PWM duty cycle until the rotorreaches a speed sufficiently high for the sensorless control to engage.However, even this solution will cause jitter during startup,particularly if there are time varying loads. Jitter can be decreased ormade imperceptible for loads with minimal initial torque or predictableinitial torque. However, some motor application/use situations (such asstarting an electric motor bike moving uphill) demand significant torquefor initiation, and the initial torque is highly unpredictable. Use ofsensorless brushless motor systems is sometimes discouraged forlow-speed high-torque maneuvers, like rock-crawling or intricate anddetailed track racing of an electric motor vehicle/bike, because in suchdifficult situations, significant jittering may occur and can lead topremature motor burnout.

FIG. 1 is a block diagram of a motor control system 10 in a three-phasepower stage, as is known in the prior art. Many three-phase motorcontrol systems 10 include a controller having a control signalgenerator 12, a gate driver 14, and a power stage 16. In case ofsensorless control, feedback circuits are also included, specifically adetection network 18 and a current sensing circuit 20, which utilizessense resistor R_(SENSE). In general, a goal of sensorless control is todetect a motor response to an applied pulse width modulated (PWM) sourcevoltage to identify rotor position and movement.

Similarly, a current sense circuit 20 may be used to detect themagnitude and direction of motor current across driven windings. Lowside shunt monitoring is used regularly. An often used configuration forlow side monitoring is shown in FIG. 1. One skilled in the art caneasily adopt alternative current sensing techniques such as monitoringphase current in each inverter branch including high-side monitoring andthis alternative technique is known to those having ordinary skill inthe art.

The control signal generator 12 is often powered from a low voltagesource. As a result, a function of the gate driver 14 includes shiftingthe low voltage control signals to levels that match input requirementsof the power stage 16. The power stage 16 includes semiconductorswitching devices. MOSFETs are shown in FIG. 1, but other devices suchas IGBTs may be used. The control signal can be made to generatetrapezoidal (a.k.a. block or 6 step commutation) or sinusoidal drivefrom the power source V_(pwr). Pulse width modulation is typically usedwith trapezoidal drive in brushless DC (BLDC) motor control. Systemsrequiring lower audible noise or lower torque ripple benefit fromsinusoidal drive.

Those skilled in the art with respect to PWM drive techniques understanda variety of modes to generate trapezoidal, sinusoidal, or othercontrol. The motor response to a PWM drive can be detected via voltageon the motor phases and/or phase current(s).

As shown in FIG. 1 for a brushless DC motor control, the power stage 16is driven such that current flows into a first motor phase (for example,phase U) and exits a second motor phase (for example, phase V). Therotor (not shown) position within the motor 30 dictates which phase pairto drive to attempt to achieve full torque and smooth (jitter-free)rotation of the rotor. The feedback controls are used to deduce rotorposition.

FIG. 2 is an illustration of a wye-connected motor 30, as is known inthe prior art. The wye-connected motor 30 in this illustration has asingle-pole pair permanent magnet rotor 32 positioned such that itssouth pole 34 is proximate to the winding of the U-phase 36. Under theseconditions, it is obvious to one skilled in the art that the W-phase 38and the V-phase 40 are the appropriate phase pair to drive in order toinitiate rotation of the rotor 32. The polarity of the permanent magnetrotor 32 determines the direction of current flow through the phase.Hence, the power stage 16 connects the W-phase 38 to V_(pwr) and theV-phase 40 to ground 24 resulting in current flow into the W-phase 38and exiting the V-phase 40, as represented with the current arrows. Anet effect of current flowing through coils W-phase 38 and V-phase 40 asshown in FIG. 2 is the formation of an electromagnet having a north poleat the W-phase 38 and a south pole at V-phase 40. This electromagnetproduces a repulsive force between permanent magnet N-pole 42 and theelectromagnet N-pole formed at the W-phase 38 and an attractive forcebetween permanent magnet N-pole 42 and the electromagnet S-pole formedat the V-phase 38.

As N- and S-poles are attracted to each other, if the electromagnetpersisted long enough in this current flow configuration, the resultingtorque will move the permanent magnet N-pole 42 to a position shortlyafter the V-phase 40 and the permanent magnet S-pole 34 to a positionshortly before the W-phase and rotation of the permanent magnet rotor 32would stop. To perpetuate rotation of the permanent magnet rotor 32, thepower stage 16 must commutate to a new phase pair. The optimumcommutation point is a function of the rotor position relative to thecoil of the undriven phase (the phase not driven by V_(pwr)). In FIG. 2,the U-phase 36 is the undriven phase. Ideally, the rotor angle wouldspan −30° to +30° with respect to alignment with the coil of theundriven phase. As this 60° span is one sixth of one electricalrevolution, it is commonly referred to as one sextant.

FIG. 3 is a 6-step commutation process further defined by Table 1, as isknown in the prior art. Given the conditions illustrated in FIG. 2, ahigh level description of the sequence of steps commonly referred to as6-step commutation process is outlined in Table 1 and furtherillustrated in FIG. 3.

TABLE 1 Six-step commutation sequence for a wye-connected motor shown inFIG. 2 Se- Driven quence Phase N-pole position S-pole position RotorStep Pair relative to phases relative to phases Angle 0 WV W + 30° to V− 30° U − 30° to U + 30° 1.25-1.75 1 WU V − 30° to V + 30° U + 30° to W− 30° 1.75-2.25 2 VU V + 30° to U − 30° W − 30° to W + 30° 2.25-2.75 3VW U − 30° to U + 30° W + 30° to V − 30° 2.75-0.25 4 UW U + 30° to W −30° V − 30° to V + 30° 0.25-0.75 5 UV W − 30° to W + 30° V + 30° to U −30° 0.75-1.25

The 6-step commutation sequence results in one electrical revolution.Given this simplified example, it is understood that a properly drivenpermanent magnet rotor will be driven one mechanical revolution whenthis six-step process is complete. An increase in number of pole pairresults in an equivalent increase in the number of electricalrevolutions per mechanical revolution. Comparing Table 1 and FIG. 2, itis understood that FIG. 2 illustrates Sequence Step 0 with the permanentmagnet N-pole 42 pushed from the W-phase 38 and pulled by the attractionto the V-phase 40. When the permanent magnet S-pole 34 reaches the U+30°position, the power stage 16 commutates to Sequence Step 1 drivingcurrent from the W-phase 38 to the U-phase 36 causing the U-phase tobecome the electromagnetic S-pole. Thus, the U-phase 36 repels or pushesthe permanent magnet S-pole 34 and the W-phase 38 attracts the S-pole,continuing the clockwise motion of the permanent magnet rotor 32.

Most current solutions to sensorless control of a brushless permanentmagnet motor utilize a symmetric pulse width modulation signal. FIG. 4Ais an illustration of one example of one symmetric pulse widthmodulation signal, as is known in the prior art. One cycle of asymmetric pulse width modulation signal may include a positive voltageV+ for a span of time T_(A) and then a negative voltage V− for the spanof time T_(B), where the absolute values of V+ and V− are equivalent anda full PWM period is T_(A)+T_(B). The span of time spent at V+ isreferenced as the energizing portion of the signal and the span of timespent at V− is referenced as the de-energized portion of the signal.FIG. 4B is an illustration of one example of one asymmetric pulse widthmodulation signal, as is known in the prior art. This signal includes aspan of time T₁ at V+ and a second span of time T₂ at approximately 0V.The sum of T₁ and T₂ represents the PWM period. Both symmetric andasymmetric driven motors are difficult to control at start-up and lowspeeds. Because of the difficulty in locating the position of the rotor,it makes it difficult to identify the commutation sequence step thatwill provide the desired torque.

Thus, a heretofore unaddressed need exists in the industry to addressthe aforementioned deficiencies and inadequacies.

SUMMARY

Embodiments of the present disclosure provide a system and method forcontrolling sequential phase switching in driving a set of statorwindings of a multi-phase sensorless brushless motor. Briefly described,in architecture, one embodiment of the system, among others, can beimplemented as follows. The system includes a controller unit having acontrol signal generator, a memory device, a processing unit, a signalacquisition device, and an analog-to-digital converter. A power stagehas a plurality of switches and receives a control signal from thecontrol signal generator and a power signal from a power source. Thepower stage drives two windings of the set of three stator windings witha multi-state pulse and leaves one stator of the three stator windingsundriven. The processing unit acquires a demodulated measured voltage onthe undriven winding. The processing unit communicates with the powerstage to change which two windings of the three stator windings aredriven when the demodulated measured voltage surpasses a threshold.

The present disclosure can also be viewed as providing a method ofcontrolling motor switching. The method includes the steps of: driving amulti-state pulse width modulated signal on two windings of a set ofthree windings; measuring a plurality of voltage values on an undrivenwinding of the set of three windings; signal processing the measuredvoltage values; and changing which two windings are driven when theprocessed voltage values exceed a commutation threshold.

Other systems, methods, features, and advantages of the presentdisclosure will be or become apparent to one with skill in the art uponexamination of the following drawings and detailed description. It isintended that all such additional systems, methods, features, andadvantages be included within this description, be within the scope ofthe present disclosure, and be protected by the accompanying claims.

BRIEF DESCRIPTION OF THE DRAWINGS

Many aspects of the disclosure can be better understood with referenceto the following drawings. The components in the drawings are notnecessarily to scale. Instead emphasis is being placed upon illustratingclearly the principles of the present disclosure. Moreover, in thedrawings, like reference numerals designate corresponding partsthroughout the several views.

FIG. 1 is a block diagram of a motor control system in a three-phasepower stage, as is known in the prior art.

FIG. 2 is an illustration of a wye-connected motor, as is known in theprior art.

FIG. 3 is a 6-step commutation process further defined by Table 1, as isknown in the prior art. FIG. 4A is an illustration of one example of onesymmetric pulse width modulation signal, as is known in the prior art.

FIG. 4B is an illustration of one example of one asymmetric pulse widthmodulation signal, as is known in the prior art.

FIG. 5 is a block diagram of a motor control system in a three-phasepower stage, in accordance with a first exemplary embodiment of thepresent disclosure.

FIG. 6 is an illustration of demodulated signals representing motorphases illustrated in FIG. 2.

FIG. 7 is an illustration of an exemplary voltage sense circuit that maybe used in conjunction with the motor control system in FIG. 5, inaccordance with the exemplary embodiment of the present disclosure.

FIG. 8 is an illustration of an exemplary current sense circuit that maybe used in conjunction with the motor control system in FIG. 5, inaccordance with the exemplary embodiment of the present disclosure.

FIG. 9 is an illustration of an exemplary multi-phase pulse widthmodulation signal for driving the motor, in accordance with an exemplaryembodiment of the present disclosure.

FIG. 10A is an illustration of the demodulated undriven phase signalassociated with the motor control system of FIG. 5, in accordance withthe exemplary embodiment of the present disclosure.

FIG. 10B is an illustration of the demodulated undriven phase signalassociated with the motor control system of FIG. 5, in accordance withthe exemplary embodiment of the present disclosure.

FIG. 11 is an illustration of a flowchart illustrating a method of usingthe motor control system 110 of FIG. 5, in accordance with the exemplaryembodiment of the present disclosure.

DETAILED DESCRIPTION

FIG. 5 is a block diagram of a motor control system 110 for athree-phase power stage 116 for a sensorless, brushless permanent magnetDC motor 30, in accordance with a first exemplary embodiment of thepresent disclosure. The motor control system 110 includes a controllerunit 160 having a control signal generator 112, a memory device 162, aprocessing unit 164, a signal acquisition device 166, and ananalog-to-digital convertor 170. The control signal generator 112 feedssix inputs into a gate driver 114. The gate driver 114, which may bepowered by an independent power source (not illustrated), controls sixMOSFET switches 168 in the power stage 116. Manipulation of the switchesdetermines current flow from the power source V_(pwr) through the statorwindings 36, 38, 40 in the motor 30.

The voltage sense circuit 118 and current sense circuit 120 are used forclosed loop control of the motor. The power stage 116 has 6 switchesgrouped in pairs. Each switch pair is configured as a half bridge. Eachswitch has a control input. The outputs of power stage 116 are fed intothe 3-phase BLDC motor windings U 36, V 40, W 38. The power stage 116 issupplied by a voltage source V_(pwr) having a DC voltage, which thepower stage uses to supply a multi-state pulse width modulation signalto the windings U 36, V 40, W 38. The current return path for thevoltage source V_(pwr) is through ground via current sense resistorR_(SENSE). The power stage 116 for a trapezoidally controlledmulti-state pulse width modulated brushless DC motor 30 typicallyenergizes two motor windings of the set of three windings 36, 38, 40 ata time.

A voltage signal is available at the undriven phase. This voltage signalcan be used to generate a commutation signal by demodulating theundriven phase voltage synchronously with the PWM switching rate. Thecommutation signal, when a near-zero drive current is present, has aperiodicity of ½ electrical revolutions. The shape of this commutationsignal is related to the action of the permanent magnet rotor 32 on thestator windings 36, 38, 40. Demodulation can be performed by simplytaking the difference in voltage between the undriven phase and theswitching in two different driven states of the PWM. When amaterially-greater-than-zero current is driven into the active pair ofterminals, the signal has an added component with a periodicity of afull electrical cycle.

FIG. 6 is an illustration of demodulated undriven phase signalsrepresenting motor phases illustrated in FIG. 2 and FIG. 3. Thesubscript D indicates the signal is from a demodulated undriven winding.Here, the undriven phase signals are illustrated for a ½ electricalcycle superimposed upon each other relative to rotor angle. The propercommutation time can be determined by monitoring the undriven phasesignal, derived from the demodulated undriven phase signal, andcommutating at a value that is a function of the motor current. As thecurrent increases, a comparison value will change, but FIG. 6 isrepresentative of a near-zero current through the driven windings.

As illustrated in FIG. 6, the dotted line U_(D) represents thedemodulated signal produced when the U-phase 36 is disconnected duringcommutation sequence step 0, and the WV phases 38, 40 are driven with aPWM wave. This drive combination is the connection that generates themost torque from the rotation position 1.25 to the 1.75 point on thex-axis, which is the sextant position

If the motor is being driven with torque pushing to the right, when 1.75point is reached, the motor is rotating in the proper direction, andcommutation from WV phases to WU phases should occur at the 1.75 point.Likewise, if the rotor is rotating counterclockwise while beingelectrically driven clockwise, such as starting an electric scooter on ahill, U_(D) has negative slope between 1.25 and 1.75. If the 1.25 pointis reached, the prior commutation phase UV or commutation sequence step5 should be switched in. These points are associated with thedemodulated signal U_(D), reaching approximately 1.5 or −1.5 volts forforward or reverse commutation respectively, illustrated as THRESHOLD inFIG. 6. When 1.75 on the x-axis is reached, upon commutating to WUphases, the demodulated signal associated with the V-phase 40, i.e.V_(D), will then be generated. If 1.25 is reached (forced in reverse),the demodulated signal associated with the W phase, i.e. W_(D), willthen be generated.

If the commutation signal component from the permanent magnets isdominant, determining the time for commutation is straightforward. Thecommutation signal from the undriven phase is derived, and whenpre-determined values are reached, the motor is advanced to the next orprior phase. The prior phase advance is important, as the load may berotating in the direction opposite to the desired rotation upon start.For maximum torque, it is important that the commutation levels berelatively accurate.

When the required starting torque is high, amaterially-greater-than-zero current is needed through the drivenwindings to generate the high torque. The commutation breakpoint isharder to determine from the undriven phase signal when the drivenwinding current is high. The commutation signal transforms substantivelywith respect to rotational position when the current has surpassed anear-zero level.

FIG. 7 is an exemplary voltage sense circuit 118 that may be used inconjunction with the motor control system 110 in FIG. 5, in accordancewith the exemplary embodiment of the present disclosure. The voltagesense circuit 118 is placed in the feedback path of a first controlloop, between the power stage outputs 116 and the controller unit signalacquisition device 166. The voltage sense circuit 118 includes aresistor network comprising resistors R1, R2, R3, R4, and R5 coupledtogether as shown in FIG. 7. Voltage sense circuit 118 has three inputsconnected to three motor terminals, U 36, V 40, W 38. The voltage sensecircuit 118 superposes motor voltage response from each phase 36, 38. 40and divides the result to level in accordance with input requirementsfrom signal acquisition 166. The result includes the voltage on theundriven phase. While similar motor control configurations includevoltage sense circuits 118, these circuits are directed to retrieving aback EMF signal and regularly filtering out the undriven phase voltageto get a cleaner back EMF.

FIG. 8 is an exemplary current sense circuit 120 that may be used inconjunction with the motor control system 110 in FIG. 5, in accordancewith the exemplary embodiment of the present disclosure. The currentsense circuit 120 is placed in the feedback path of a second controlloop, between a current sense resistor R_(SENSE) and the controller unitsignal acquisition device 166. The power supply voltage levels ofcurrent sense circuit 120 and controller unit 160 are approximately thesame. Current sense circuit 120 includes an amplifier 174 configured fordifferential measurement of voltage across R_(SENSE), as shown in FIG.5. The amplifier 174 input common-mode voltage and gain are set suchthat amplifier output is at approximately mid-supply to facilitatemonitoring of R_(SENSE) current flowing in positive and negativedirection.

As previously described herein, the power stage 116 supplies amulti-stage pulse width modulation signal to the windings. Themulti-stage pulse width modulation signal may be a signal that canswitch between an asymmetric pulse width modulation signal and asymmetric pulse width modulation signal. The multi-stage pulse widthmodulation signal may also be a signal that has three voltage levels ina single period rather than the two voltage levels offered by thesymmetric and asymmetric signals.

FIG. 9 is an illustration of an exemplary multi-phase pulse widthmodulation signal for driving the motor, in accordance with an exemplaryembodiment of the present disclosure. The multi-phase pulse widthmodulation signal includes a positive voltage V+ for T₁, a negativevoltage V− for T₂, and a zero voltage for T₃ such that the period of thesignal is T₁+T₂+T₃. In comparison to the signals shown in FIG. 4A andFIG. 4B, the positive voltage stage in FIG. 9 is comparable to theenergized stage in each of FIG. 4A and FIG. 4B. The negative stage inFIG. 9 is comparable to the de-energized stage in FIG. 4A. The zerostage in FIG. 9 is comparable to the de-energized stage in FIG. 4B.These three stages can be achieved with the same power stage 116switching that was used to obtain the comparable stages in FIG. 4A andFIG. 4B.

Using this three stage pulse width modulation signal may increase signalto noise ratio (SNR), allowing for more effective use of the demodulatedmeasured voltage to control commutating driven windings. A duration ofeach of the positive voltage portion, negative voltage portion and azero voltage portion may be calculated based on operating conditions andmotor characteristics and the duration of each portion may be alteredfor consecutive cycles to adapt to operating conditions and motorcharacteristics. The period of the multi-state pulse width modulatedsignal may be changed for subsequent cycles.

The motor control system 110 may be used to control a motor 30, such asthe motor 30 illustrated in FIG. 2. FIG. 10A is an illustration of thedemodulated undriven phase signal associated with the motor controlsystem 110, in accordance with the exemplary embodiment of the presentdisclosure. Signals V and W are driven signals on two terminals of themotor 30. Initially, FIG. 10A illustrates a proximate 50% duty cycle PWMwith complementary drive. The drive phase voltage will normally be avalue between ground and the power supply voltage. The PWM signal thenswitches to an asymmetric drive. Signals V and W remain the drivenwindings on the motor 30. The W+ gate and the V− gate will close whenenergizing while the other four gates are open. While de-energizing, theW− gate and the V− gate will close, disconnecting the set of windingsfrom power source V_(pwr) and connecting the W and V phases to eachother and ground. The drive phase voltage will remain a value betweenground and the power supply voltage. The PWM signal closes out byreturning to the initial proximate 50% duty cycle PWM with complimentarydrive.

Typical switching frequencies are in the range of 1 kHz to 25 KHz,depending on motor size and construction as well as other factors. Thesignal at the undriven phase is shown in FIG. 10A as signal U. Signal Uchanges as a function of rotor position which varies the magnetic fieldsin the stator. The demodulated undriven phase signal U_(D), which isused for position sensing, is derived by measuring the voltagedifference on signal U between the high b_(n) and low a_(n) level whilethe pulse with modulation signal is symmetric. This voltage differencecan be viewed as demodulation of the position signal from the PWMsignal. When the PWM signal is asymmetric, the demodulated undrivenphase signal U_(D) is derived by measuring the voltage during theenergizing phase with respect to a reference voltage. This comparison ofthe measured voltage with a reference voltage is at least part of thedemodulation step. The demodulated signal is compared with anestablished threshold, such as the threshold shown in FIG. 6, and usedto determine the commutation breakpoint where the power stage outputwill switch to a next winding pair to drive. The illustration of U_(D)in FIG. 10A is analogous to the 1.25-1.75 rotor angle portion of theU_(D) curve in FIG. 6 operating with steady rotor movement.

FIG. 10B is an illustration of the demodulated undriven phase signalassociated with the motor control system of FIG. 5, in accordance withthe exemplary embodiment of the present disclosure. The signal at theundriven phase is shown in FIG. 10A as signal U. Signal U changes as afunction of rotor position which varies the magnetic fields in thestator. The demodulated undriven phase signal U_(D), which is used forposition sensing, is derived by measuring the voltage difference onsignal U between the high b_(n) and low a_(n) level for the three-statepulse with modulation signal. This voltage difference can be viewed asdemodulation of the position signal from the PWM signal. The demodulatedsignal is compared with an established threshold, such as the thresholdshown in FIG. 6, and used to determine the commutation breakpoint wherethe power stage output will switch to a next winding pair to drive. Theillustration of U_(D) in FIG. 10A is analogous to the 1.25-1.75 rotorangle portion of the U_(D) curve in FIG. 6 operating with steady rotormovement.

FIG. 11 is an illustration of a flowchart 200 illustrating a method ofusing the motor control system 110 of FIG. 5, in accordance with theexemplary embodiment of the present disclosure. It should be noted thatany process descriptions or blocks in flow charts should be understoodas representing modules, segments, portions of code, or steps thatinclude one or more instructions for implementing specific logicalfunctions in the process, and alternate implementations are includedwithin the scope of the present disclosure in which functions may beexecuted out of order from that shown or discussed, includingsubstantially concurrently or in reverse order, depending on thefunctionality involved, as would be understood by those reasonablyskilled in the art of the present disclosure.

As is shown by block 202, a multi-state pulse width modulated signal isdriven on two windings of a set of three windings. A voltage of anundriven winding of the set of three windings is measured (block 204).The measured voltage is demodulated (block 206). A different pair ofwindings of the set of three windings is driven when the demodulatedmeasured voltage exceeds a threshold (block 208).

The step of changing which two windings are driven may involve changingwhich phases are driven after the demodulated measured voltage hasexceeded the threshold for a set period of time. The undriven voltagesignal may experience noise, and that noise may cause the threshold tobe surpassed prematurely and temporarily. Verifying that the demodulatedmeasured voltage continues to exceed the threshold for a period of timediminishes the possibility that the threshold is surpassed as a resultof noise instead of properly identified rotor position.

When the multi-state pulse width modulated signal changes between anasymmetric and symmetric drive mode, as illustrated in FIG. 10A, can bebased on a number of factors. The output on the undriven winding hasdifferent noise issues for low current and high current in asymmetricand symmetric modes. To enhance signal to noise ratio (SNR), the drivepulse may be changed between symmetric and asymmetric when a currentthrough the two windings surpasses a signal switching threshold. Thedrive signal may be changed between symmetric and asymmetric when thecommutation signal is captured. In another exemplary use, the controllerunit 160 may have a clock that tracks how long a current commutationsequence step has been active and change between symmetric andasymmetric drive modes when a timing threshold is passed.

The system may switch between symmetric and asymmetric drive modes whena commutation threshold is proximate to the demodulated undriven voltagevalues. Both forward and reverse commutation thresholds may beconsidered when determining proximity of the thresholds.

The threshold may be set as a function of the pulse width modulatedsignal. For instance, as an amplitude of the pulse width modulationsignal increases, the absolute value of the thresholds should increaseto properly compensate for the undriven winding voltage also increasingin value. The threshold may be predetermined and modified as a functionof a characteristic of the pulse width modulated signal. Similarly, thedemodulated measured voltage value may be modified within the motorcontroller as a function of the pulse width modulated signal to allowthe demodulated measured voltage value to intersect the threshold at theproper rotor rotation angles. The demodulated measured voltage may bemodified by scaling the demodulated measured voltage.

While the pulse width modulation signal can be useful to project ways tomodify the thresholds or the demodulated measured voltage, another valuethat can be useful is the current over the driven windings. The motorcontroller can use the current sense circuit to identify the currentvalue over the driven windings. The demodulated measured voltage can bemodified as a function of the current through the driven windings. Thethreshold can be modified as a function of the current through thedriven windings.

First Exemplary Commutation Breakpoint Calculation

A pulse width modulation signal is provided to two windings at a levelthat provides a near zero average current (I_(min)) over the twowindings. A first set of voltage data representing the motor voltageresponse signal on the undriven phase 36, spanning at least an entiresextant, is obtained. A first set of current data representing thedriven phase current is collected corresponding to each data point inthe first set of undriven voltage data. The process is repeated with apulse width modulation signal that provides a mid-level drive phasecurrent (a.k.a. I_(mid)) and again with a pulse width modulation signalthat provides an approximately maximum drive phase current (a.k.a.I_(max)).

A first set of coefficients representing the influence of mid-levelvalues of current is calculated based on first and second current datasets.Coeff_(midCurrent)=(V _(MTR)(I _(mid))−V _(MTR)(I _(min)))/(I _(mid) −I_(min))

Where V_(MTR) is the demodulated motor voltage response signal based onthe undriven phase 36.

A second set of coefficients representing the influence of max-levelvalues of current is calculated based on first and third current datasetsCoeff_(maxCurrent)=(V _(MTR)(I _(max))−V _(MTR)(I _(min)))/(I _(max) −I_(min))

The effect of current on the commutation signal is different in oddsextants compared to even sextants. Therefore, said first and secondsets of coefficients are created for both even and odd sextants.

Coeff_(midCurrent)(odd)

Coeff_(midCurrent)(even)

Coeff_(maxCurrent)(odd)

Coeff_(maxCurrent)(even)

The resultant coefficient values can be used as-is under specificconditions. For example, if an application runs at specific currentsbecause the motor drives known loads, then the coefficients can bestored in a lookup table. At each operating current level, thecoefficients can then be read from the table and used to compensate theundriven phase signal for that current.

Another method of modifying the threshold and/or demodulated voltageincludes transforming the resultant coefficient values into slope andintercept values for even and odd sextants, which can then be generallyapplied for a wide set of current values. The Slope and Intercept valuesare stored in memory.

The coefficient as a function of current is calculated as:Coefficient(I)=slope*I _(avg)+intercept

In this equation, I_(avg) is the average driven phase current, obtainedin this example via amplifier 174 in difference configuration monitoringlow side shunt resistor and generally described as current sense blockin FIG. 5 and FIG. 8. The amplifier output is sampled and digitized inboth the on and off portions of the PWM cycle. The values are digitallyprocessed to produce the average motor phase current in the PWM cycle.The Slope and Intercept values may be obtained from memory device 162.Sextant parity determines whether slope and intercept data for odd oreven sextants is used.

Slope is effectively calculated as ΔV/ΔI, hence, Coefficient(I) hasunits of resistance.

A correction factor as a function of current is then calculated as:V _(CF)(I)=I _(avg)*Coefficient(I)

Controller unit memory device 162 contains constant values representingmotor characteristics. Constant value(s) for commutation breakpoint isstored in memory device 162. Slope and intercept values are stored inmemory device 162.

Processing unit 164 performs arithmetic calculations based on stored andmeasured data. Specifically, the correction factor, V_(CF)(I), iscalculated and the motor voltage response on the undriven phase isdemodulated. The processing unit 164 inverts the polarity of thedemodulated signal in every other sextant such that the slope of thedemodulated signal with respect to the direction of the applied torqueis positively independent of the sextant. The processing unit 164modifies the demodulated signal with the correction factor in accordancewith the winding current. The processing unit 164 calculates directionof the demodulated signal based on its slope between commutationbreakpoints, thereby confirming direction of rotation. A differencebetween first and second demodulated signal data points taken betweenconsecutive commutation breakpoints is compared to a threshold value. Adifference value greater than the threshold value indicates positiveslope, while a difference value less than the threshold value indicatesnegative slope. The definition of slope by way of comparison to athreshold value is arbitrary. For example, a difference value less thana threshold value could just as well define a positive slope.

The processing unit 164 compares a modified/corrected demodulated signalto a stored forward commutation breakpoint. At least one occurrence ofthe combination of a modified demodulated signal having value greaterthan the forward commutation breakpoint value and confirmed forwarddirection of rotation results in processing unit 164 controlling thecontrol signal 112 to commutate the power stage 116 to a next phasepair. Requiring multiple occurrences of the satisfying condition priorto commutating may increase system robustness. The processing unit 164compares a modified/corrected demodulated signal to a stored reversecommutation breakpoint. At least one occurrence of the combination of amodified demodulated signal having value less than the reversecommutation breakpoint value and confirmed reverse direction of rotationresults in processing unit 164 controlling PWM 112 to commutate thepower stage 116 to a previous phase pair. Requiring multiple occurrencesof the satisfying condition prior to commutating may increase systemrobustness.

An average current across the driven windings can be acquired a numberof ways, including measurement and modeling, some of which are known tothose skilled in the art. One useful method for obtaining the currentacross the driven windings is averaging a current measured by an analogto digital convertor and a current sense mechanism. As is discussedabove, the average current is used to modify at least one of thethresholds and the demodulated measured voltage.

When the rotor rotates fast enough, relative to other motorcharacteristics and operating conditions, a reliable back EMF signalbecomes available. Use of a reliable back EMF signal to controlcommutation from driven pair to driven pair is well known in the art.Thus, the techniques disclosed herein are designed for controllingcommutation when the rotor is not moving or is rotating at speeds belowwhich a reliable back EMF signal is available. The motor controlswitches to the back EMF commutation technique when a rotational speedof the rotor surpasses a speed threshold such that the reliable back EMFsignal is available.

It should be emphasized that the above-described embodiments of thepresent disclosure, particularly, any “preferred” embodiments, aremerely possible examples of implementations, merely set forth for aclear understanding of the principles of the disclosed system andmethod. Many variations and modifications may be made to theabove-described embodiments of the disclosure without departingsubstantially from the spirit and principles of the disclosure. All suchmodifications and variations are intended to be included herein withinthe scope of this disclosure and protected by the following claims.

What is claimed is:
 1. A method of controlling motor switching, themethod comprising: manipulating a gate controller with a control unit todrive a multi-state pulse width modulated signal on two windings of aset of three windings within a motor; measuring a plurality of voltagevalues with a voltage sense circuit on an undriven winding of the set ofthree windings within a motor; demodulating the measured voltage values;and changing which two windings are driven when the demodulated voltagevalues exceed a commutation threshold with the controller unitmanipulating the gate controller.
 2. The method of claim 1, wherein themulti-state pulse width modulated signal further comprises changing froma first pulse width modulated signal to a second pulse width modulatedsignal driving the two windings of the set of three windings, whereinthe first pulse width modulated signal operates in a symmetric mode andthe second pulse width modulated signal operates in an asymmetric mode.3. The method of claim 2, wherein changing from the first pulse widthmodulated signal to the second pulse width modulated signal is performedwhen a current through the two windings surpasses a signal switchingthreshold.
 4. The method of claim 2, wherein changing from the firstpulse width modulated signal to the second pulse width modulated signalis performed when an energizing portion of the first pulse widthmodulated signal exceeds a timing threshold.
 5. The method of claim 2,wherein changing from the first pulse width modulated signal to thesecond pulse width modulated signal is performed after changing whichtwo windings are driven.
 6. The method of claim 1, wherein themulti-state pulse width modulated signal further comprises changing froma first pulse width modulated signal to a second pulse width modulatedsignal driving the two windings of the set of three windings, whereinthe first pulse width modulated signal operates in an asymmetric modeand the second pulse width modulated signal operates in a symmetricmode.
 7. The method of claim 6, wherein changing from the first pulsewidth modulated signal to the second pulse width modulated signal isperformed when a current through the two windings falls below a signalswitching threshold.
 8. The method of claim 6, wherein changing from thefirst pulse width modulated signal to the second pulse width modulatedsignal is performed when an energizing portion of the first pulse widthmodulated signal is below a timing threshold.
 9. The method of claim 6,wherein changing from the first pulse width modulated signal to thesecond pulse width modulated signal is performed when the demodulatedvoltage values are proximate to the commutation threshold.
 10. Themethod of claim 1, wherein the multi-state pulse width modulated signalfurther comprises a hybrid cycle that includes a positive voltageportion, a negative voltage portion, and a zero voltage portion withinone cycle.
 11. The method of claim 10, wherein a duration of each of thepositive voltage portion, negative voltage portion, and a zero voltageportion is calculated based on operating conditions and motorcharacteristics.
 12. The method of claim 1, wherein a period of themulti-state pulse width modulated signal is changed for a subsequentcycle.
 13. A system for controlling motor switching, the systemcomprising: a controller unit comprising a control signal generator, amemory device, a processing unit, a signal acquisition device, and ananalog-to-digital converter; a power stage having a plurality ofswitches in communication with the control signal generator, wherein thepower stage receives a control signal from the control signal generatorand a power signal from a power source, wherein the power stage drivestwo windings of the set of three stator windings with a multi-statepulse and leaves one stator of the three stator windings undriven;wherein the processing unit device unit in communication with the signalacquisition device acquires a demodulated measured voltage on theundriven winding; and wherein the processing unit in communication withthe power stage through the control signal generator and incommunication with the memory device communicates with the power stageto change which two windings of the three stator windings are drivenwhen the demodulated measured voltage surpasses a threshold stored onthe memory device.
 14. The system of claim 13, wherein the power stagedrives the two windings with a symmetric drive signal and an asymmetricdrive signal during a single commutation sequence step.
 15. The systemof claim 13, wherein the processor determines whether the power stagedrives the two windings with a symmetric drive signal or an asymmetricdrive signal, wherein the processor determination is based on an outputfrom the signal acquisition device.
 16. The system of claim 13, whereina single multi-state pulse period includes three states, that comprisesa positive voltage, a negative voltage, and a ground.
 17. The system ofclaim 13, further comprising a clock in the controller unit, wherein theclock is programmed to measure time elapsed within a current commutationstep and the multi-state pulse further comprises the power stageswitching between a first pulse width modulation drive signal and asecond pulse width modulation drive signal when a determined time haselapsed within the current commutation step.
 18. The system of claim 13,wherein the multi-state pulse further comprises the power stageswitching between a first pulse width modulation drive signal and asecond pulse width modulation drive signal when the demodulated measuredvoltage is proximate to the threshold.
 19. The system of claim 13,wherein the processing unit demodulates the measured voltage.